In-band and out-of-band signal detection for automatic gain calibration systems

ABSTRACT

An embodiment of the present invention provides an automatic gain control system for a wireless receiver that quickly differentiates desired in-band signals from high power out-of-band signals that overlap into the target band. The system measures power before and after passing a received signal through a pair of finite impulse response filters that largely restrict the signal&#39;s power to that which is in-band. By comparing the in-band energy of the received signal after filtering to the total signal energy prior to filtering, it is possible to determine whether a new in-band signal has arrived. The presence of this new in-band signal is then verified by a multi-threshold comparison of the normalized self-correlation to verify the presence of a new, desired in-band signal.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention is directed to communication systems. Moreparticularly, the invention is directed to receivers in wirelesscommunication systems. Even more particularly, the present invention isdirected to automatic gain control systems for such wirelesscommunication system receivers.

2. Background of the Related Art

The use of receivers in wireless systems such as radio and cellularcommunication systems is well-known in the art. FIG. 1 shows a typicalsuperheterodyne receiver design 10. Here, a radio frequency (RF) signalis received on antenna 15 and provided to RF amplifier 20. The RF signalis amplified by the RF amplifier 20 and in mixer 25 mixed with a signalfrom a local oscillator 30. This produces an intermediate frequency (IF)signal that is amplified in an IF amplifier 35 and filtered in abandpass filter 40. The filtered IF signal is again amplified by an IFamplifier 45 and mixed in a product detector 50 with a signal from abeat frequency oscillator 55. The result is a signal that is amplifiedby a baseband amplifier 60 and digitized for further processing by ananalog-to-digital (A/D) converter 65.

In such receivers, less amplifier gain is needed for strong signals, andit is important that a very strong signal not be amplified to the pointthat when amplified it distorts received information signals, overloadssystem components and possibly damages the components. For this reason,receivers typically have some sort of automatic gain control (AGC)system which controls one or more of the system amplifiers 20, 35, 45and 60 to maintain the amplified signals within certain ranges (thiscontrol may be, e.g., through a bias applied to the amplifiers). In FIG.1, the AGC unit 70 receives an IF input output by the IF amplifier 45and uses it to generate bias signals controlling the RF amplifier 20 andthe IF amplifiers 35 and 45.

SUMMARY OF THE INVENTION

An embodiment of the present invention provides an automatic gaincontrol system for a wireless receiver that quickly differentiatesdesired in-band signals from high power out-of-band signals that overlapinto the target band. The system measures power before and after passinga received signal through a number of filters that largely restrict thesignal's power to that which is in-band. By comparing the in-band energyof the received signal after filtering to the total signal energy priorto filtering, it is possible to determine whether a new in-band signalhas arrived. The presence of this new in-band signal is then verified bya multi-threshold comparison of the normalized self-correlation toverify the presence of a new, desired in-band signal.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other aspects of an embodiment of the present invention arebetter understood by reading the following detailed description of thepreferred embodiment, taken in conjunction with the accompanyingdrawings, in which:

FIG. 1 shows the structure of a communications receiver of the priorart;

FIG. 2 shows the structure of a communications receiver of an embodimentof the present invention;

FIG. 3 shows the structure of an automatic gain control mechanism in theembodiment of FIG. 2;

FIGS. 4 and 5 show desired characteristics of a gain-controlled signal;

FIG. 6 shows characteristics of A/D converter saturation in theembodiment; and

FIGS. 7A, 7B, 8A and 8B show characteristics of in-band and out-of-bandsignals.

DETAILED DESCRIPTION OF PRESENTLY PREFERRED EXEMPLARY EMBODIMENTS

The basic structure of a receiver of an embodiment of the presentinvention is shown in FIG. 2. Here, a wideband antenna 115 receives aradio frequency (RF) RF signal and provides it to an RF amplifier 120,and a particular channel or signal within the band is preferablyselected by varying the local oscillators 130 and 180. In theembodiment, the RF signal preferably conforms to the IEEE 802.11astandard, has a frequency in the 5 GHz band and is quadrature modulatedto carry 6 to 54 Mbps. In this embodiment, the signal can carry up to 54Mbits of data and lies within one of twelve 20 MHz wide slots, eightwithin a 5.15-5.35 GHz band and four within a 5.75-5.85 GHz band. Thesignal in this embodiment is a coded orthogonal frequency divisionmultiplexed (OFDM) signal using 52 subcarriers spaced 312.5 kHz apart.It is understood, however, that while the following detailed descriptionof the present invention is made in the context of an IEEE 802.11asystem, that the inventions described herein have application to manydifferent types of communication systems, and are not limited to systemsoperating within the IEEE 802.11a standard. For example, as describedhereinafter the present invention described operating upon the short andlong training symbols in an IEEE 802.11a system, but it is noted thatthe teachings related thereto can be generalized to any trainingsequence made up of one or more sinusoids. Thus, for example, powermeasurements can be made based upon half of a period of a slowestfrequency sinusoid that exists within a training symbol containing aplurality of sinusoids, with each of the plurality of sinusoids having afrequency that is an integer multiple of the slowest frequency sinusoid.In 802.11a system this translates to half of a short training symbolsequence.

The amplified RF signal is mixed with a signal from a local RFoscillator 130 supplied to an RF mixer 125 to generate an intermediatefrequency (IF) signal that is fed to an IF amplifier 135. Preferably,the sum of the frequencies of the local RF oscillator 130 and local IFoscillator 180 are in the range 5.15-5.35 and 5.75-5.85 GHz, with theratio of the RF oscillator frequency to the IF oscillator frequencybeing 4:1. In the embodiment, the local oscillators 130 and 180 arepreferably in a floating IF arrangement in which they both are variable,rather than a fixed IF arrangement where, e.g., only the RF localoscillator 130 is variable.

The amplified IF signals are supplied to an in-phase mixer 175-IP and aquadrature mixer 175-Q, respectively. One of the in-phase mixer 175-IPand the quadrature mixer 175-Q is directly driven by a local IFoscillator 180, and the other of the in-phase mixer 175-IP and thequadrature mixer 175-Q is driven by the local IF oscillator signal afterit is phase-shifted by 90° in a phase shifter 185. In this way, in-phase(IP) and quadrature (Q) components of the received RF signal areobtained at the outputs of the in-phase mixer 175-IP and quadraturemixer 175-Q, respectively.

The mixed IF signals pass through low-pass filters 140-IP and 140-Q toselect the desired channel and remove spectrally distant components notof interest, and are amplified by two series of baseband amplifiers145-IP and 145-Q. Though two baseband amplifiers are shown in eachbranch, a different number of amplifiers may be used. Almost any desiredbaseband gain step arrangement may be developed using basebandamplifiers having appropriately selected programmable gains in aparticular order.

Preferably, the low-pass filters 140-IP and 140-Q are two-poleelliptical filters having a 3 dB corner at 28 MHz. Moving from theanalog to digital domain, the baseband amplifier outputs are fed to A/Dconverters 190-IP and 190-Q which digitize the in-phase and quadraturecomponent signals, preferably with a frequency of 80 MHz, to aresolution of nine bits, and an input dynamic range of −500 mV to 500mV.

Preferably, the A/D converters are pipeline A/D converters; however, theinvention is not so limited. For example, sigma-delta or otherconverters may be used in their place.

An analog channel filter and/or anti-aliasing filter may advantageouslybe placed before the A/D converters 190-IP and 190-Q. In the preferredembodiment, the combination of the analog filters perform adjacentblocker rejection of 4 dB and an alternate blocker rejection of 20 dB.With a worst case of an adjacent blocker 16 dB larger and an alternateblocker 32 dB larger, a received blocker at the A/D converter input canbe 12 dB higher than the in-band signal. As is known in the art, anadjacent blocker is an interference signal adjacent to or overlappingthe frequency band of interest, while an alternate blocker is aninterference signal farther away from the frequency band of interest.

The digitized I/Q component signals are provided to an automatic gaincontrol (AGC) unit 170 whose operation with respect to the presentinvention will be described in greater detail below. The AGC 170analyzes the I/Q component signals as described in greater detail belowand generates gain control signals based thereon. These gain controlsignals are provided to the amplifiers 120, 135, 145-IP and 145-Q asshown by the dotted line in FIG. 2.

More specifically, as shown in FIG. 3 the digitized IF signals from theA/D converters 190-IP and 190-Q are passed through leaky bucket filters245-IP and 245-Q and finite impulse response (FIR) filters 205-IP,210-IP and 205-Q, 210-Q. The first FIRs 205-IP and 205-Q are decimationfilters that eliminate every other sample from their respective streamsto reduce the data sampling rate from 80 MHz to 40 MHz for a normal 8.5MHz single-sided bandwidth packet. The second FIRs 210-IP and 205-Q arestandard low-pass filters which remove any residual adjacent or aliasedblockers before sending the data to the self-correlator 225 and a powerdetector 220. Two power measurements are taken within the AGC 170—onefrom the output of the second FIR filter 210 by the power detector 220,and another from the output of A/D converters 190-IP and 190-Q byanother power detector 215. These measurements are provided to AGCcontrol logic 230 as will be described in greater detail below.

Although this embodiment uses digital FIRs, other types of filters,including analog filters, may be used in their place. If the system isnot oversampled, the filters are preferably analog.

AGC control logic 230 receives the power measurements from powerdetectors 215 and 220 and uses them to control a gain control generator235 to output analog gain control signals for each of the RF amplifier120, the IF amplifier 135, and individual ones of the basebandamplifiers 145-IP and 145-Q. In the embodiment, the AGC control logic230 provides a control word, ten bits in length in the preferredembodiment, to the gain control generator 235, and the gain controlgenerator 235 generates appropriate control signals for the amplifiers.These gain control signals are fed back to the RF amplifier 120, the IFamplifier 135 and the baseband amplifiers 145-IP and 145-Q to controlthe gain of each as described above.

The embodiment uses an application-specific integrated circuit toimplement the AGC control logic 230; however, an appropriatelyprogrammed processor, either embedded or discrete, or other appropriatedevice, may be used as well.

It should be noted that although FIG. 3 shows various components withinthe AGC 170 to be separate from one another, it is possible that two ormore units may be integrated into one. For example, the AGC controllogic 230 is shown separately from the FIRs 205, 210, power detectors215, 220 and self-correlation unit 225; however, several of these may becombined into a single processor appropriately programmed to performthese functions. Further, a programmed processor need not be used andone or more of these components can be implemented in dedicatedhardware.

The AGC 170 may control a DC offset control unit 240 to provide analogoffset control signals to one or more of the baseband amplifiers 145-IPand 145-Q. DC offset control is done to ensure that the analog signalsprovided to the amplifiers and A/D converters 190-IP and 190-Q areproperly centered and quantized.

AGC Operation

In the embodiment, the control logic 230 first checks to see if thesignal is sufficiently saturating either of the A/D converters 190-IPand 190-Q. If so, a quick drop gain control procedure is executed; ifnot, a base gain control procedure, also described below, is executed.

Next, the AGC base gain control logic 230 determines whether thereceived signal is within a preferred range as described below. If so,no gain control is needed; otherwise, a gain control procedure describedin greater detail below is executed.

Then, the AGC system 170 attempts to identify an in-band signal usingstrong signal and weak signal detection techniques, as described ingreater detail below. If a signal is found, the detection process iscomplete; if not, the detection process is repeated on the next portionof the signal. Weak signal detection and strong signal detection areindependent and complementary features. As described further herein, forstrong signal detection, it is determined that a signal may exist by thearrival of a stronger signal necessitating a drop in receive gain,whereas for weak signal detection, it is determined that a signal mayexist due to a sudden increase in measured in-band power at leastproportional to the increase in total power at the ACC (while notrequiring a gain change), followed shortly by a self-correlationexceeding thresholds. It is noted that it is preferable to disable weaksignal detection, typically for a few microseconds, if a gain change ismade, since self-correlation will not be valid until the entire viewingwindow for self-correlation is filled with post-gain-change values.Thus, weak-signal detection is used for arriving signals not largeenough relative to blockers or noise to cause gain changes, and strongsignal detection for larger arriving signals. And for strong signaldetection, that a new signal has arrived is determined based uponwhether a coarse gain drop or quickdrop gain results, as describedbelow.

AGC Base Gain Control for a Coarse Gain Change

In operation, the AGC 170 must adjust receiver gains so that thereceived signal can properly be quantized by the A/D converter 190. Ifthis signal is too big at the A/D converter input, the signal will bedistorted by saturation. If the signal is too small at the A/D converterinput, the quantization noise of the A/D converter 190 will render thereceived signal-to-noise (S/N) ratio too low for correct detection. Forthis purpose, the AGC control logic 230 digitally controls the analogvariable gain stages mentioned above using the gain control unit 235.Preferably, the embodiment's gain control has a dynamic range of 93dB-51 dB in the combined RF and IF stages 120 and 135 and 42 dB in thebaseband stage 145.

The power detector 215 estimates the total digitized power at the A/Dconverters 190-IP and 190-Q by summing a window of instantaneous powercalculations for half of a preamble short symbol window in an 802.11asignal (400 ns) for a total of 16 samples. For example, consider asignal coming out of a nine-bit A/D converter 190 with a range of [−256,255], and measurement of power for this signal over a 16-bit samplewindow in half a preamble short symbol window. To do this, the AGCcontrol logic 230 calculates the instantaneous power adcpwr1 on the A/Dconverter output stream adcoutput as $\begin{matrix}{{adcpwr1} = {{\sum\limits_{k = 0}^{15}\quad \left( {{real}\left( {{adcoutput}\quad\lbrack k\rbrack} \right)} \right)^{2}} + \left( {{imag}\left( {{adcoutput}\quad\lbrack k\rbrack} \right)} \right)^{2}}} & (1)\end{matrix}$

This power measurement is then put into a log table, where its maximumvalue is zero. Thus, for a fully railed output with every value at −256,the logarithmic table output would be zero. The power of a full-railsinusoid would be −3 dB; if every sample were 128, the power would be −6dB, etc.

The AGC control logic 230 uses this total power estimate to keep thesignal in-range at the A/D converters 190-IP and 190-Q. If the signalpower is determined to be out of range (but not saturating the A/Dconverters 190-IP and 190-Q), a coarse gain change will be made to putthe signal back in range. More specifically, if AGC control logic 230detects the total measured power adcpwr1 (in the embodiment, within therange −63-0 dB) is greater than the maximum desired A/D converter signalsize, the desired gain value gaintarget, which is a signal size that isset large enough so that quantization noise is small enough, but alsosmall enough that ADC saturation is not an issue, including the size ofthe signal and any potential blocker, is reduced in a course gain dropby the AGC control logic 230 of the equation

gaintarget=gaintarget+(coarsepwr _(—) const−adcpwr1)  (2)

where coarsepwr_const is an additional gain for coarse gain drop (FIG.4), e.g., −17 dB. This additional gain loss is used because the incomingsignal may be too large to quantize but not large enough to trigger aquick drop as described in greater detail below—for example, if thesignal saturates occasionally but not enough to trigger a quick drop. Insuch cases, it is useful to drop the gain by more than the gaintargetvalue indicates, based on power measurements of a saturated waveform—avery aggressive drop. Thus, the empirically determined coarsepwr_constvalue is added to increase the gain drop to more quickly converge on thedesired signal size. The result is used to generate appropriate controlsignals for the amplifiers via the gain control generator 235.

If the total measured power adcpwr1 is less than the minimum desired A/Dconverter signal size, the desired gain value gaintarget is increased bythe AGC control logic 230 of the equation

gaintarget=gaintarget+(totalsizedesired−adcpwr1)  (3)

where totalsizedesired is the target A/D converter signal size duringcoarse gain changes, i.e., the desired size of the A/D converter outputin the absence of a desired signal (FIG. 5)—about −17 dB in thepreferred embodiment.

AGC Quick Drop Gain Control

If the received signal is saturating the A/D converters 190-IP and 190-Qoften, a precise power measurement may not be obtained; however, it iscertain that the signal is well out of range. This information can beused to quickly reduce the gain. More specifically, a saturation counteradcsat is established by the AGC control logic 230 to count the numberof saturations of either the I or Q A/D converter output samples. A pairof saturation thresholds adcsat_thrh and adcsat_thrl, which can bechanged by downloading a different threshold, are used to counter anypossible lack of A/D converter range. Thus, a saturation will bedetected if

adcoutput≧(adcsat _(—) thrh+192)  (4)

or if

adcoutput≦(adesat _(—) thrl−256)  (5)

where adcsat_thrh is a high threshold less than the maximum A/Dconverter output value which designates saturation on the high side ofthe A/D converter output, adcsat_thrl is a low threshold value greaterthan the minimum A/D converter output value which designates saturationon the low side of the A/D converter output (FIG. 6) and the constantvalues are implementation-dependent. adcsat_thrh is set to be slightlyless than the maximum A/D converter output, while adcsat_thrl is set tobe slightly higher than the minimum A/D converter output. This is usefulbecause it allows signals that are close to saturation of, but do notactually saturate, the A/D converter to be classified as saturationsignals for more flexibility. If the number of saturations of the A/Dconverter output samples during a sample window of adcsat_icount cycles(preferably less than or equal to eight, the number of cycles in thequarter-symbol 802.11a measurement window) exceeds a saturationthreshold amount adcsat_thresh a quick gain drop is instructed by theAGC control logic 230, and gaintarget is reduced by a predeterminedamount quick_drop, e.g., a −30 dB change in gain. In the preferredembodiment, the adcsat_thresh is set for at a threshold of 12saturations in an 8-cycle window (with saturations independentlypossible on I and Q ADCs).

This technique may advantageously be implemented in the following way.After calibration or any gain change, an AGC settling time occurs. Afterthat adcpwr1, the variable corresponding to the amount of measuredpower, is reset and an acc_count counter, preferably an eight-bitincremental counter cycling continuously during AGC operation, also isreset.

The following events will happen of the counter acc_count:

mod(acc_count, 16)=0: reset adcpwr1 accumulator

mod(acc_count, 16)=1: clear reset on adcpwr1 accumulator

mod(acc_count, 16)=2: store adcpwr1

mod(acc_count, 16)≦adcsat_icount and adcsat=1 (asserted when set_thresh

saturations, e.g. eight saturations, have been counted), the saturationcounter has exceeded adcsat_thresh and a quick gain drop should beexecuted. As shown, in the preferred embodiment, the adcpwr1 values arecomputed every 16 cycles, and the system looks for adcsat to be assertedprior to the first 4 bits of the counter registering a value greaterthan adcsat_icount (preferably 8).

AGC Packet Detection

Once the received signal is in-range, the AGC control logic 230 detectsthe presence of a desired packet. For this purpose, the AGC controllogic 230 determines an in-band power estimation, uses the FIR filters205-IP, 210-IP and 205-Q and 210-Q to reduce all adjacent and alternateblockers to 20 dB below the in-band signal power at 802.11a specifiedmaximum levels, and compares adcpwr1 and firpwr1 as describedhereinafter. This is done to obtain information about whether quantizedsignal energy at the A/D converter 190-IP or 190-Q is in-band orout-of-band—information which helps in finding the desired packets.

More specifically, consider the signal shown in FIG. 7A. Calculating thepower of an A/D converter output as described above might determine thatit has an overall power of, say, −12 dBr, where dBr is a measure of theRMS size of signals below the full rail signal size described above withreference to Equation (1). Passing through the second FIR 210-IP or210-Q as shown in FIG. 7B, however, the signal loses most of its powerand is reduced to a level of about −25 dBr—a decrease of roughly 85%.Since most of the signal's power was blocked by the bandpass FIR 210-IPor 210-Q, it is presumed to be an out-of-band signal.

Referring to the signal shown in FIG. 8A, this signal too has an overallpower of about −12 BDr as measured at the A/D converter output. Passingthrough the second FIR 210-IP or 210-Q as shown in FIG. 8B, however,only reduces its power to approximately −15 dBr—a decrease of only about10%. Since most of the signal's power was passed by the bandpass FIR210-IP or 210-Q, it is presumed to be an in-band signal.

With this understanding, the in-band power is calculated as the sum ofinstantaneous power measurements, preferably in a 32 sample, 0.8 μswindow similar to the overall power calculation adcpwr1 described above.firpwr1 is the power based on the lowest of some number of samples thatis less than the entire number of samples obtained, such as 28 out of 32samples in the 32 sample window in detector 220. It is noted that thenumber of samples for firpwr1 is greater than the number of samples foradcpwr1 because firpwr1 is being used for fine gain control, whereprecision is important, whereas adcpwr1 is being used for coarse gainchanges, where a slightly noisy power estimate will do. It is also notedthat while for purposes of this in-band power calculation less than theentire number of samples is preferably used, that other post analog todigital converter processing that takes place using such samples willtypically use all the samples obtained.

This less than the entire number of samples is used because duringperiods of interference, e.g., at symbol boundaries of the interferers,a temporary in-band power spike may occur due to high-frequencycomponents of interferers at the symbol transition becoming in-bandcomponents in the desired band. This will artificially show up as a stepin the in-band power. Windowing at the transmitter of the interferer,e.g., using a value which is half the previous value added to half thesubsequent value at the symbol boundary, reduces this somewhat, as doeslowpass filtering, so that the aggregate spectrum passes the necessaryspectral mask. These instantaneous high frequency peaks, althoughlowered, can still exist. When an adjacent interferer is present, thistemporary high frequency component in the interferer is actually in-bandfor the desired signal, so that the in-band power measurement when nodesired signal is present can get a quick spike for a few samples,looking like an increase in the in-band power. To combat this, thelowest 28 of the 32 samples are used so this temporary spike is nulledout by not counting those values, and thresholds are adjustedaccordingly to compensate for the reduced power measurement due to themissing four samples. Once a signal of interest is present, however, allsamples are preferably used in creating the power measurement to make adetailed measurement. This second power measurement is calledfirpwr_all. Using the power information described above, desired signalscan be found in two ways: strong signal detection and weak signaldetection. Strong signal detection will be described first.

Strong Signal Detection

Any time a coarse gain drop or quick gain drop as described aboveoccurs, a flag strongsignal is set by the AGC control logic 230. Thisflag remains high until the signal is determined to be in range at theA/D converter 190-IP or 190-Q, and the algorithm proceeds to make afirpwr1 measurement as described above. At this point, flag_relpwr iscalculated as

flag_(—) relpwr=set if (firpwr1>relpwr+adcpwr)  (6)

(where relpwr is an empirical thresholding variable related to theabsolute digital size of the in-band signal relative to the absolutetotal digital signal at the A/D converter 190-IP or 190-Q), thusattempting to see that most of the computed power is in-band. Ifflag_relpwr is high and strongsignal is high, then a new, very strongin-band signal has appeared. In this way, the embodiment permitsexamination of an oversampled incoming signal having digitizedfrequencies beyond a desired frequency range due to oversampling, anddetermine whether most of its power is in-band before determining that adesired signal has been found.

Thus, when flag_relpwr is high and strongsignal is high, thesignal_found flag is asserted, a fine gain change is made as describedbelow and the AGC process is completed once the number of consecutivegain changes is equal to or greater than the minimum number of gainchanges deemed to constitute a successful AGC operation, i.e., whenthere have been enough gain changes to ensure a full programmableamplifier ramp-up when the system is turned on.

Weak Signal Detection

In weak signal detection, the normalized self-correlation of shortsequences as defined below is measured to look for anything in-band witha periodicity of 0.8 μs in the preferred embodiment. This is a two-stepprocess performed concurrently with the above-described strong signaldetection process. First, the system waits for the normalizedself-correlation as measured by the self-correlation processor 225 toexceed a first normalized self-correlation magnitude threshold valuem1thres.

The self-correlation processor 225 preferably measures self-correlationof 802.11a packets by taking 32 samples in a short training symbol atthe beginning of a packet and comparing each of the samples to acorresponding sample from the preceding short training symbol. Morespecifically, the self-correlation of an A/D converter stream adcoutputis given by $\begin{matrix}{{self\_ corr} = \frac{\left\lbrack {\Sigma \quad {{{adcoutput}\lbrack n\rbrack} \cdot {{conj}\left( {{adcoutput}\left\lbrack {n - 32} \right\rbrack} \right)}}} \right\rbrack^{2}}{\Sigma \quad {{adcoutput}\lbrack n\rbrack}^{2}}} & (7)\end{matrix}$

where the denominator is a normalization factor. One can see that thenumerator will be relatively high when x[n] and x[n−32] are identicaland relatively low when, e.g., they are uncorrelated. Thus, this measurecan serve as a good indicator of self-correlation.

Detecting when the self-correlation output exceeds m1thres can thusdetect the existence of an incoming packet; however, it would alsodetect interferers, since they can have structures that can alsoself-correlate. For this reason, the embodiment advantageously employsanother test. Once the normalized self-correlation exceeds ml thres, thesystem enters a loop and for m1count_max cycles counts in a variable mltally the number of times the normalized self-correlation exceeds asecond normalized self-correlation magnitude threshold value m2thres,where m2thres is less than or equal to m1thres. If m1tally>m2count_thr,a threshold of the count of normalized self-correlation>m2thres, beforem1count_max (a window length for the self-correlation count) cycles haveelapsed, weak signal detection may be detected.

As noted above, the windowing technique based on m1count_max is usedbecause both interferers and noise may have a self-correlation thatmomentarily exceeds a threshold, but the chances of this occurringdiminish when windows of samples obtained over consecutive periods oftime are used. For example, a subsequent window will contain many of thesame samples as the previous window, but the previous window will notcontain the most recent sample from the subsequent window, and thesubsequent window will not contain the oldest sample from the previouswindow. Thus, for example, if two 802.11a symbols in adjacent channelsare sent, such that they are separated in frequency by 20 MHz, the last0.8 μs of the first symbol will exactly match the next 0.8 μs guardperiod of the next symbol, creating self-correlation, but this spikewill rapidly fade, in comparison with a preamble where a flat normalizedself-correlation result is expected for the preamble duration.

Thus, the embodiment provides a way of performing a two-thresholdwindowing process on a self-correlation measurement. One threshold isused to determine that a signal may be present in-band, and the numberof times a second threshold is exceeded in different windows of offsetsamples is counted to further determine if that in-band signal is adesired signal. This is done to combat temporary correlation of thermalnoise as well as to combat self-correlation during the data segment ofan interferer.

Additionally, for further robustness against thermal noise andinterferers, the embodiment preferably requires that to enable a weaksignal detection result, a potential detected packet must increase thein-band signal power by at least a certain amount and that the increasebe at least proportional to any increase in the total signal power, thesignal power being of at least a certain minimum size. This providesextra sensitivity when a new in-band signal comes in below an interfereror near the noise floor, thus not triggering strong signal detection butworthy of a look for weak signal detection.

At least three things may stop weak signal detection from occurring oncem1tally>m1thresh. First, if ycOK=0, weak signal detection will notoccur. ycOK is a decrementing counter that is reset to ycOKmax (in theembodiment, four) to enable weak signal detection if it is determinedthat an increase in the in-band signal of a certain size (flag_firstep)and at least proportional to any increase in the total power (flagrelstep) with the measured firpwr1 of at least a certain minimum size(flag_firpwr), then it is possible that a new in-band signal has come inbelow an interferer or near the noise floor, thus not triggering strongsignal detection but worthy of a look for weak signal detection. Toensure that such recognition occurs within a limited period of time, theabove must happen while ycOK>0 if it happens at all. To perform thesestep calculations, old values of firpwr1 and adcpwr are stored asfirpwr{2-4} and adcpwr{2-4}. Enough values are stored so that if thesignal is detected during a programmable amplifier ramp, enoughdifference will exist between the first and last measurements to exceedthe given threshold.

Another reason why weak signal detection might not occur is becausegc_count is greater than zero. gc_count measures the time since the lastgain change in short symbol increments, getting decremented by the AGCcontrol logic 230 for every valid firpwr1 measurement from its startingvalue of three after a gain change. The idea is that after a gainchange, there is a minimum amount of time until a self-correlation isvalid.

Finally, weak signal detection will not occur if the signal has alreadybeen found with another method, since then there is no need to find itusing weak signal detection.

AGC Packet Detection—DC Offset Elimination

The above double threshold arrangement is successful in reducing falsepacket detects on interferers during weak signal detection; however, itis not particularly successful in preventing false detection withrespect to DC signals, which always self-correlate. There is typically asmall DC component at the output of the A/D converter 190, so theembodiment uses a two-tap DC notch filter as a leaky bucket filter—morespecifically, a two-tap IIR filter having a transfer curve of the form$\begin{matrix}{{y\lbrack n\rbrack} = {{\frac{\alpha - 1}{\alpha}{y\left\lbrack {n - 1} \right\rbrack}} + {\frac{1}{\alpha}{x\lbrack n\rbrack}}}} & (8)\end{matrix}$

where x is the input signal, y is the output signal and α is a filterparameter (in this case, 32)—which uses an estimate of the DC levelprovided by the AGC logic control 230 to cancel the DC component out.The AGC control logic 230 obtains this level from a lookup table basedon current gain settings.

AGC Completion Processing

Once the signal is found via either strong signal detection or weaksignal detection, fine gain changes will be made, in the preferredembodiment if consec_gainchanges<min_gainchanges. And in the preferredembodiment, every fine gain change will be made based upon the equation

gain_change=adc_desired_size−firpwr1_all  (9)

consec_gainchanges begins at zero for strong signal detection and twofor weak signal detection, since it is meant to be a coarse measure oftime spent in the AGC, and it takes approximately two gain change timesto perform a windowed self-correlation. It is incremented every coarseand fine gain change. It is reset when no gain change is made and strongsignal detection does not decide that a signal is present. This featureis meant to ensure that a minimum amount of time is spent in the AGC,for either more precise gain or to be sure the gain is set after the PAis done ramping.

The preferred embodiments described above have been presented forpurposes of explanation only, and the present invention should not beconstrued to be so limited. Variations on the present invention willbecome readily apparent to those skilled in the art after reading thisdescription, and the present invention and appended claims are intendedto encompass such variations as well.

What is claimed is:
 1. A method of estimating whether a received strongsignal is an in-band signal based upon power estimates, the methodcomprising the steps of: inputting the received strong signal;estimating a power of the received strong signal by summinginstantaneous power calculations; providing the received strong signalto a filter section; using the filter section to pass frequencycomponents of the received strong signal within a desired band offrequencies and at a desired data sampling rate to obtain a filteredsignal; estimating a power of the filtered signal by summinginstantaneous power calculations; and comparing the power estimate ofthe received strong signal to the power estimate of the filtered signalto obtain an indication that the received signal is estimated as beingin-band.
 2. The method of claim 1, wherein estimating the power of thereceived strong signal and estimating the power of the filtered signalare performed by summing instantaneous power calculations within windowsof the received strong signal and the filtered signal, respectively. 3.The method of claim 2, wherein extreme instantaneous power calculationsin the power estimate of the received strong signal and in the powerestimate of the filtered signal are not used in the respectiveestimates.
 4. The method of claim 3, wherein highest magnitudeinstantaneous power calculations in the power estimate of the receivedstrong signal and in the power estimate of the filtered signal are notused in the respective estimates.
 5. The method of claim 4 wherein priorto the step of estimating the power of the received strong signal thereis included the step of removing DC offset from the input signal.
 6. Themethod of claim 4 wherein the step of estimating the power of thereceived strong signal uses a different number of signal instantaneouspower calculations than the step of estimating the power of the filteredsignal.
 7. The method of claim 4 wherein the received strong signalpower estimate is made based upon less than a received whole shorttraining symbol.
 8. The method of claim 4 wherein the received strongsignal power estimate is made based on half of a period of a slowestfrequency sinusoid that exists within a training symbol containing aplurality of sinusoids, with each of the plurality of sinusoids having afrequency that is an integer multiple of the slowest frequency sinusoid.9. The method of claim 2 wherein prior to the step of estimating thepower of the received strong signal there is included the step ofremoving DC offset from the input signal.
 10. The method of claim 2wherein the step of estimating the power of the received strong signaluses a different number of signal instantaneous power calculations thanthe step of estimating the power of the filtered signal.
 11. The methodof claim 1, wherein comparing the power estimates of the received strongsignal and the filtered signal includes determining whether the filteredsignal power estimate is at least a certain portion of the receivedsignal power estimate.
 12. The method of claim 1 wherein prior to thestep of estimating the power of the received strong signal there isincluded the step of removing DC offset from the input signal.
 13. Themethod of claim 1, further including the step of determining that thereceived strong signal is a desired signal after the indication has beenprovided.
 14. The method of claim 1 wherein the received signal powerestimate is performed on less than a whole short training symbol. 15.The method of claim 14 wherein the received signal power estimate isperformed on about one half of a received whole short training symbol.16. A method of estimating whether an input signal is an in-band signal,the method comprising the steps of: inputting the input signal;converting the input signal to a received digital signal using an analogto digital converter; determining whether the received digital signal issaturating the analog to digital converter and if so identifying thedigital signal as a strong signal type; estimating the power of thereceived digital signal by summing instantaneous power calculations;determining whether the power of the received digital signal is greaterthan a desired level and if so identifying the received digital signalas the strong signal type; providing the received digital signal to afilter section; using the filter section to pass frequency components ofthe received digital signal within a desired band of frequencies toobtain a filtered signal; estimating a power of the filtered signal bysumming instantaneous power calculations; comparing the power estimateof the received digital signal to the power estimate of the filtereddigital signal to obtain an indication that the received digital signalis an in-band signal type; and determining that the received digitalsignal is an in-band signal if the received digital signal is indicatedas being of the in-band signal type and is identified as being of thestrong signal type.
 17. The method of claim 16 further including thesteps of: reducing a gain provided by an automatic gain control circuitto an automatic gain control amplification section if saturation isdetermined; and reducing a gain provided by the automatic gain controlcircuit to the automatic gain control amplification section if the powerof the received digital signal is greater than the desired level. 18.The method of claim 16, wherein estimating the power of the receiveddigital signal and estimating the power of the filtered signal areperformed by summing instantaneous power calculations within windows ofthe received digital signal and the filtered signal, respectively. 19.The method of claim 18, wherein extreme instantaneous power calculationsin the power estimate of the received digital signal and in the powerestimate of the filtered signal are not used in the respectiveestimates.
 20. The method of claim 19, wherein highest magnitudeinstantaneous power calculations in the power estimate of the receiveddigital signal and in the power estimate of the filtered signal are notused in the respective estimates.
 21. The method of claim 20 whereinprior to the step of estimating the power of the received digital signalthere is included the step of removing DC offset from the input signal.22. The method of claim 20 wherein the step of estimating the power ofthe received digital signal uses a different number of signalinstantaneous power calculations than the step of estimating the powerof the filtered signal.
 23. The method of claim 20 wherein the receiveddigital signal power estimate is performed on less than a received wholeshort training symbol.
 24. The method of claim 23 wherein the receiveddigital signal power estimate is performed on about one half of areceived whole short training symbol.
 25. The method of claim 18 whereinprior to the step of estimating the power of the received digital signalthere is included the step of removing DC offset from the input signal.26. The method of claim 18 wherein the step of estimating the power ofthe received digital signal uses a different number of signalinstantaneous power calculations than the step of estimating the powerof the filtered signal.
 27. The method of claim 18, wherein comparingthe power estimates of the received strong signal and the filteredsignal includes determining whether the filtered signal power estimateis at least a certain portion of the received signal power estimate. 28.The method of claim 16 wherein prior to the step of estimating the powerof the received strong signal there is included the step of removing DCoffset from the input signal.
 29. The method of claim 16 wherein thereceived signal power estimate is determined base upon less than a wholeshort training symbol.
 30. The method of claim 29 wherein the receivedsignal power estimate is determined based upon about one half of areceived whole short training symbol.
 31. The method of claim 16 furtherincluding the step of making a fine automatic gain control adjustment ifit was determined that an in-band signal exists.
 32. A method ofdetermining that a received digital signal of a strong signal typeexists within an input signal including the steps of: converting theinput signal to the received digital signal using an analog to digitalconverter; estimating the power of the received digital signal bysumming instantaneous power calculations; determining whether the powerof the received digital signal is greater than a desired level and if soidentifying the received digital signal as the strong signal type; anddetermining whether the received digital signal saturates the analog todigital converter and if so identifying the received digital signal asthe strong signal type, wherein the step of determining includes:counting a number of times that a plurality of samples of the receiveddigital signal exceeded the predetermined saturation threshold; andcomparing the number of times with a predetermined saturation countthreshold, such that when the predetermined saturation threshold isexceeded, the received digital signal is determined to saturate theanalog to digital converter.
 33. The method of claim 32 furtherincluding the step of reducing a gain provided by an automatic gaincontrol circuit to an automatic gain control amplification section ifthe power of the received digital signal is greater than the desiredlevel.
 34. The method of claim 33 wherein the step of reducing the gainreduces the gain by an amount that is based upon the estimated power anda predetermined gain target value.
 35. The method of claim 32 furtherincluding the step of: reducing a gain provided by an automatic gaincontrol circuit to an automatic gain control amplification section ifthe received digital signal is determined to saturate the analog todigital converter.
 36. The method of claim 35 wherein the step ofreducing the gain reduces the gain a predetermined amount.
 37. A systemfor detecting in-band signals, the system comprising: a filter sectionfor filtering a received digital signal to pass frequency components ofthe received strong signal within a desired band of frequencies and at adesired data sampling rate to obtain a filtered signal; a power detectorfor estimating a power of the filtered signal and for estimating a powerof the received digital signal by summing instantaneous powercalculations of the respective signals; and control logic for, based onthe estimated powers of the received digital signal and the filteredsignal, respectively, determining whether the received signal is anin-band signal.
 38. The system of claim 37, wherein the filter sectionincludes a finite impulse response filter.
 39. The system of claim 37,wherein the filter section includes a decimation filter.
 40. The systemof claim 37, wherein the filter section includes a low-pass filter. 41.The system of claim 37, wherein the power detector includes separatepower detectors for estimating the powers of the received digital signaland the filtered signal, respectively.
 42. The system of claim 41,wherein each of the power detectors includes a sampler configured totake instantaneous power calculation samples of its respective signalwithin windows thereof.
 43. The system of claim 41 wherein each of thepower detectors does not use highest magnitude instantaneous powercalculations in the power estimate of its respective signal.
 44. Thesystem of claim 37, wherein the filtered signal power estimate uses atleast certain signal instantaneous power calculations from the receivedsignal power estimate.
 45. The system of claim 37 further including a DCoffset control unit for removing DC offset from the input signal. 46.The system of claim 37 wherein the control logic includes: means fordetermining whether the power of the received digital signal is greaterthan a desired level and if so identifying the received digital signalas the strong signal type.
 47. The system of claim 46 further including:means for reducing a gain provided by an automatic gain control circuitto an automatic gain control amplification section if the power of thereceived digital signal is greater than the desired level.
 48. Thesystem of claim 47 wherein the means for reducing the gain reduces thegain by an amount that is based upon the estimated received digitalsignal power and a predetermined gain target value if the power of thereceived digital signal is greater than the desired level.
 49. Thesystem of claim 47 further including: means for determining whether thereceived digital signal saturates the analog to digital converter and ifso identifying the digital signal as the strong signal type.
 50. Thesystem of claim 49 wherein the means for determining includes means forcounting a number of times that a plurality of samples of the receiveddigital signal exceeded a predetermined saturation threshold; and meansfor comparing the number of times with a predetermined saturation countthreshold, such that when the predetermined saturation threshold isexceeded, the received digital signal is determined to saturate theanalog to digital converter.
 51. The system of claim 49 furtherincluding: means for reducing the gain provided by the automatic gaincontrol circuit to the automatic gain control amplification section ifthe received digital signal is determined to saturate the analog todigital converter.
 52. The method of claim 47 wherein the means forreducing the gain reduces the gain a predetermined amount if thereceived digital signal is determined to saturate the analog to digitalconverter.
 53. A method of determining that a received digital signal ofthe strong signal type exists within an input signal including the stepsof: converting the input signal to a received digital signal using ananalog to digital converter; determining whether the received digitalsignal saturates the analog to digital converter, the step ofdetermining including: counting a number of times that a plurality ofsamples of the received digital signal exceeded the predeterminedsaturation threshold; and comparing the number of times with apredetermined saturation count threshold, such that when thepredetermined saturation threshold is exceeded, the received digitalsignal is determined to saturate the analog to digital converter; andreducing a gain provided by an automatic gain control circuit to anautomatic gain control amplification section if the received digitalsignal is determined to saturate the analog to digital converter. 54.The method of claim 53 wherein the step of reducing the gain reduces thegain a predetermined amount.